1. Field of the Invention
This invention generally relates to the field of power converters, and more particularly, to full bridge converters that operate with zero-voltage switching (ZVS).
2. Description of the Prior Art
Full-bridge switching power converters are a common DC/DC conversion topology used for supplying high power to a wide variety of electronic components, including telecommunication equipments, computers, servers, etc. A DC/DC switching converter contains an input port typically coupled to a DC power source, such as a battery, to provide DC power to a load connected to an output port. In general, full-bridge converters utilize two bridge legs with four switching elements in the primary side of a transformer that isolates the input port from the output port. Usually, a pulse width modulator is used to regulate the output voltage under various load conditions.
Important considerations in the design of converters include power density and conversion efficiency. Power density relates to the amount of delivered power relative to the volume occupied by the power converter, for example in terms of cubic centimeter or cubic inch. As such, for a specified power output, higher density power converters require a smaller size than lower density power converters. Another parameter influencing size of the converter is volt-second product that is applied to inductive converter elements. The volt-second product is a measure of maximum voltage applied during a period of time to the terminals of the inductive elements.
Power efficiency is the ratio of input power to output power, usually expressed in terms of percentages. Power efficiency is a measure of internal power losses when converting input power to output power. Switching power converters usually suffer from two types of losses: conduction losses and switching losses. Conduction losses are associated with energy dissipation in the form of heat due to resistive converter elements. Switching losses are associated with the switching elements of the power converter. A common approach utilized to minimize switching losses is known as zero voltage switching (ZVS).
A conventional full-bridge (FB) ZVS pulse-width-modulated (PWM) converter used in high-power applications is shown in FIG. 1. The major features of this converter are constant-frequency operation and ZVS of the primary switches with a reduced circulating energy. The control of the output voltage at a constant frequency is achieved by a phase-shift technique. In this technique, the turn-on of a switch in the Q3–Q4 leg of the bridge is delayed, i.e., phase shifted, with respect to the turn-on instant of the corresponding switch in the Q1–Q2 leg. If there is no phase-shift between the legs of the bridge, no voltage is applied across the primary of the transformer and, consequently, the output voltage is zero. On the other hand, if the phase shift is 180°, the maximum volt-second product is applied across the primary winding, which produces the maximum output voltage.
ZVS is achieved by discharging the energy stored injunction capacitances C1, C2, C3 and C4, of the switches Q1, Q2, Q3 and Q4 into an inductive element in order to avoid hard switching conditions, before the switches are turned on. In the circuit of FIG. 1, the ZVS of the lagging-leg switches Q3 and Q4 is achieved primarily by discharging the corresponding junction capacitances C3 and C4 via an output filter inductor LF. Since the inductance of LF is relatively large, the storable energy in LF is sufficient to discharge this junction capacitances C3 and C4 of the switches Q3 and Q4 and, consequently, to achieve ZVS even at very light load currents, for example, when the delivered power to the load is substantially less than the rated power of the converter. However, the discharge of the junction capacitances C1 and C2 of the leading-leg switches Q1 and Q2 is done by the energy stored in the leakage inductance LLK of the transformer because during the switching of Q1 or Q2 the transformer primary is shorted by the simultaneous conduction of rectifiers DR1 and DR2 that carry the output filter inductor current. Since the leakage inductance LLK is small, the energy stored in LLK is also small so that ZVS of Q1 and Q2 is hard to achieve even at relatively high load currents, for example, when the delivered power to the load is at the rated power of the converter. The ZVS range of the leading-leg switches can be extended to lower load currents by increasing the leakage inductance of the transformer and/or by adding a large external inductor on the primary side in series with the primary of the transformer, as shown in FIG. 2. If properly sized, the external inductor LP can store enough energy to achieve ZVS of the leading-leg switches even at light load currents. However, a large external inductor also stores an excessive amount of energy at full load, i.e., when the delivered power is equal to the rated power, which produces a relatively large circulating energy that increases semiconductor components stress, as well as conversion efficiency.
In addition, a large inductance in series with the primary of the transformer extends the time that is needed for the primary current to change direction from positive to negative, and vice versa. This extended commutation time results in a loss of duty cycle on the secondary of the transformer, which further decreases the conversion efficiency. In order to provide full power at the output, the secondary-side duty-cycle loss can be compensated by reducing the turn-ratio of the transformer. With a smaller transformer turn-ratio, however, the reflected output current into the primary is increased, which increases the primary-side conduction losses, thereby creating undesired heat. Moreover, a smaller turn-ratio of the transformer increases the voltage stress on the secondary-side rectifiers. As a result, rectifiers with a higher voltage rating that typically have higher conduction losses may be required.
Yet another limitation of the circuit in FIG. 1 is a severe parasitic ringing at the secondary of the transformer during the turn-off of the rectifier. This ringing is caused by the resonance of the rectifier's junction capacitance with the leakage inductance of the transformer and the external inductor, if any. In order to control this secondary-side ringing, a snubber or clamp circuit can be employed. For implementations without an external inductor in the primary of the transformer, a passive R-C-D snubber on the secondary-side is usually used, as indicated by dashed lines in FIG. 2. For implementations with an external primary inductor the more efficient way to control the secondary-side ringing is to use primary-side clamp diodes DC1 and DC2, as shown in FIG. 2. This primary-side clamp circuit consisting of two clamp diodes was introduced and its operation described in U.S. Pat. No. 5,198,969 by Redl and Balogh. While this circuit offers a practical and efficient solution to the secondary-side ringing problem, it does not offer any improvement in the secondary-side duty-cycle loss.
A FB ZVS-PWM converter that achieves ZVS of the primary switches in the entire load and input voltage range with virtually no loss of secondary-side duty cycle and with minimum circulating energy was describe in U.S. Pat. No. 6,356,462 by Jang and Jovanović, which is assigned to the assigner of the present invention. This converter, shown in FIG. 3, employs a primary-side inductor connected to the primary center tap of the power transformer to achieve a wide-range ZVS. This converter utilizes the energy stored in the inductor to discharge the capacitance across any switch before it is turned on for achieving ZVS. At high loads, when the phase shift of the bridge is decreased, the inductor is subjected to the lowest volt-second product, with a minimum of 0 volts. At light loads, when the phase shift of the bridge is increased, the inductor is subjected to the highest volt-second product, with a maximum of +−VIN/2. As described in U.S. Pat. No. 6,356,462, since the energy stored in the inductor is decoupled from the output, the inductor does not participate in the parasitic resonance with the junction-capacitance of the secondary-side rectifier.
By properly selecting the value of the inductance of the inductor, the primary switches in the converter of FIG. 3 can achieve ZVS even at no load. This is because the energy required to create ZVS conditions at light loads does not need to be stored in the leakage inductance, and thus, the transformer leakage inductance can be minimized. As a result, the loss of duty cycle on the secondary-side is also minimized, which maximizes the turns ratio of the transformer and, consequently, minimizes the conduction losses. In addition, the minimized leakage inductance of the transformer significantly reduces the secondary-side ringing caused by the resonance between the leakage inductance and junction capacitance of the rectifier, which greatly reduces the power dissipation of a snubber circuit that is used to dampen the ringing.
The energy storage capacity of the inductor used for storage of energy used to discharge junction capacitances that achieve ZVS is determined based on e=L*i2, where e, L, and I correspond to energy, inductance and current, respectively. As described above, the conventional techniques have relied on increasing the inductance of the inductor used for ZVS energy storage. However, with the ever increasing need to increase power density or otherwise reduce the size of power converters, there is a requirement to decrease the size of the ZVS storage inductor. More specifically, a converter is needed that offers ZVS switching in a wide load range with an acceptable duty cycle loss and reduced primary-side inductor size that is also capable of storing the required energy for achieving ZVS.